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50 MHz to 9 GHz 65 dB TruPwr Detector ADL5902
FEATURES
Accurate rms-to-dc conversion from 50 MHz to 9 GHz Single-ended input dynamic range of 65 dB No balun or external input matching required Waveform and modulation independent, such as GSM/CDMA/W-CDMA/TD-SCDMA/WiMAX/LTE Linear-in-decibels output, scaled 53 mV/dB Transfer function ripple: <0.1 dB Temperature stability: <0.3 dB All functions temperature and supply stable Operates from 4.5 V to 5.5 V from -40C to +125C Power-down capability to 1.5 mW Pin-compatible with the 50 dB dynamic range AD8363
FUNCTIONAL BLOCK DIAGRAM
VPOS
3
POS
10
ADL5902
TEMPERATURE SENSOR
8
TEMP VSET
7
INHI 14
X2
IDET
INLO 15
LINEAR-IN-dB VGA (NEGATIVE SLOPE) NC
2
X2
ITGT G=5
6
VOUT
NC 16
BIAS AND POWERDOWN CONTROL
VREF 2.3V
5
Power amplifier linearization/control loops Transmitter power controls Transmitter signal strength indication (TSSI) RF instrumentation
1
11
12
9
4
TADJ/PWDN
VREF
VTGT
COMM
COMM
Figure 1.
GENERAL DESCRIPTION
The ADL5902 is a true rms responding power detector that has a 65 dB measurement range when driven with a single-ended 50 source. This feature makes the ADL5902 frequency versatile by eliminating the need for a balun or any other form of external input tuning for operation up to 9 GHz. The ADL5902 provides a solution in a variety of high frequency systems requiring an accurate measurement of signal power. Requiring only a single supply of 5 V and a few capacitors, it is easy to use and capable of being driven single-ended or with a balun for differential input drive. The ADL5902 can operate from 50 MHz to 9 GHz and can accept inputs from -62 dBm to at least +3 dBm with large crest factors, such as GSM, CDMA, W-CDMA, TD-SCDMA, WiMAX, and LTE modulated signals. The ADL5902 can determine the true power of a high frequency signal having a complex low frequency modulation envelope or can be used as a simple low frequency rms voltmeter. Used as a power measurement device, VOUT is connected to VSET. The output is then proportional to the logarithm of the rms value of the input. In other words, the reading is presented directly in decibels and is scaled 1.06 V per decade, or 53 mV/dB; other slopes are easily arranged. In controller mode, the voltage applied to VSET determines the power level required at the input to null the deviation from the set point. The output buffer can provide high load currents. The ADL5902 has 1.5 mW power consumption when powered down by a logic high applied to the PWDN pin. It powers up within approximately 5 s to its nominal operating current of 73 mA at 25C. The ADL5902 is supplied in a 4 mm x 4 mm, 16-lead LFCSP for operation over the wide temperature range of -40C to +125C. The ADL5902 is also pin-compatible with the AD8363, 50 dB dynamic range TruPwrTM detector. This feature allows the designer to create one circuit layout for projects requiring different dynamic ranges. A fully populated RoHS-compliant evaluation board is available.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c)2010 Analog Devices, Inc. All rights reserved.
08218-001
APPLICATIONS
NC 13
CLPF
26pF
ADL5902 TABLE OF CONTENTS
Features .............................................................................................. 1 Applications ....................................................................................... 1 Functional Block Diagram .............................................................. 1 General Description ......................................................................... 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 Absolute Maximum Ratings............................................................ 7 ESD Caution .................................................................................. 7 Pin Configuration and Function Descriptions ............................. 8 Typical Performance Characteristics ............................................. 9 Theory of Operation ...................................................................... 15 Square Law Detector and Amplitude Target .............................. 15 RF Input Interface ...................................................................... 16 Small Signal Loop Response ..................................................... 16 Temperature Sensor Interface ................................................... 17 VREF Interface ........................................................................... 17 Temperature Compensation Interface ..................................... 17 Power-Down Interface ............................................................... 18 VSET Interface ............................................................................ 18 Output Interface ......................................................................... 18 VTGT Interface .......................................................................... 18 Basis for Error Calculations ...................................................... 18 Measurement Mode Basic Connections.................................. 19 Setting VTADJ.................................................................................. 20 Setting VTGT .................................................................................. 20 Choosing a Value for CLPF ............................................................ 20 Output Voltage Scaling .............................................................. 22 System Calibration and Error Calculation.............................. 23 High Frequency Performance................................................... 23 Low Frequency Performance .................................................... 24 Description of Characterization ............................................... 24 Evaluation Board Schematics and Artwork ................................ 25 Assembly Drawings .................................................................... 26 Outline Dimensions ....................................................................... 27 Ordering Guide .......................................................................... 27
REVISION HISTORY
4/10--Revision 0: Initial Version
Rev. 0 | Page 2 of 28
ADL5902 SPECIFICATIONS
VS = 5 V, TA = 25C, ZO = 50 , single-ended input drive, RT = 60.4 , VOUT connected to VSET, VTGT = 0.8 V, CLPF = 0.1 F. Negative current values imply that the ADL5902 is sourcing current out of the indicated pin. Table 1.
Parameter Test Conditions Min Typ Max Unit
OVERALL FUNCTION Frequency Range RF INPUT INTERFACE Input Impedance Common Mode Voltage 100 MHz 1.0 dB Dynamic Range Maximum Input Level, 1.0 dB Minimum Input Level, 1.0 dB Deviation vs. Temperature
50 to 9000 Pins INHI, INLO, ac-coupled Single-ended drive, 50 MHz 2000 2.5 63 3 -60 -0.11/+0.25 -0.22/+0.15 -0.35/+0.25 -0.22/+0.15 53.8 -62.1 61 1 -60 +0.3/-0.2 -0.1/0 +0.3/-0.4 -0.1/0 53.7 -62.8
MHz V dB dBm dBm dB dB dB dB mV/dB dBm
CW input, TA = +25C, VTADJ = 0.5 V Calibration at -60 dBm, -45 dBm, and 0 dBm Calibration at -60 dBm, -45 dBm, and 0 dBm Deviation from output at 25C -40C < TA < +85C; PIN = 0 dBm -40C < TA < +85C; PIN = -45 dBm -40C < TA < +125C; PIN = 0 dBm -40C < TA < +125C; PIN = -45 dBm
Logarithmic Slope Logarithmic Intercept 700 MHz 1.0 dB Dynamic Range Maximum Input Level, 1.0 dB Minimum Input Level, 1.0 dB Deviation vs. Temperature
-45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm CW input, TA = +25C,VTADJ = 0.4 V Calibration at -60 dBm, -45 dBm, and 0 dBm Calibration at -60 dBm, -45 dBm, and 0 dBm Deviation from output at 25C -40C < TA < +85C; PIN = 0 dBm -40C < TA < +85C; PIN = -45 dBm -40C < TA < +125C; PIN = 0 dBm -40C < TA < +125C; PIN = -45 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm CW input, TA = +25C, VTADJ = 0.4 V Calibration at -60 dBm, -45 dBm, and 0 dBm Calibration at -60 dBm, -45 dBm, and 0 dBm Deviation from output at 25C -40C < TA < +85C; PIN = 0 dBm -40C < TA < +85C; PIN = -45 dBm -40C < TA < +125C; PIN = 0 dBm -40C < TA < +125C; PIN = -45 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm
dB dBm dBm dB dB dB dB mV/dB dBm
Logarithmic Slope Logarithmic Intercept 900 MHz 1.0 dB Dynamic Range Maximum Input Level, 1.0 dB Minimum Input Level, 1.0 dB Deviation vs. Temperature
61 1 -60 +0.3/-0.2 0/-0.1 +0.3/-0.4 0/-0.1 53.7 -62.7
dB dBm dBm dB dB dB dB mV/dB dBm
Logarithmic Slope Logarithmic Intercept
Rev. 0 | Page 3 of 28
ADL5902
Parameter Test Conditions Min Typ Max Unit
Deviation from CW Response
11.02 dB peak-to-rms ratio (CDMA2000) 5.13 dB peak-to-rms ratio (16 QAM) 2.76 dB peak-to-rms ratio (QPSK) CW input, TA = +25C, VTADJ = 0.4 V Calibration at -60 dBm, -45 dBm, and 0 dBm Calibration at -60 dBm, -45 dBm, and 0 dBm Deviation from output at 25C -40C < TA < +85C; PIN = 0 dBm -40C < TA < +85C; PIN = -45 dBm -40C < TA < +125C; PIN = 0 dBm -40C < TA < +125C; PIN = -45 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm, and 0 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm CW input, TA = +25C, VTADJ = 0.4 V Calibration at -60 dBm, -45 dBm, and 0 dBm Calibration at -60 dBm, -45 dBm, and 0 dBm Deviation from output at 25C -40C < TA < +85C; PIN = 0 dBm -40C < TA < +85C; PIN = -45 dBm -40C < TA < +125C; PIN = 0 dBm -40C < TA < +125C; PIN = -45 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm 12.16 dB peak-to-rms ratio (four-carrier W-CDMA) 11.58 dB peak-to-rms ratio (LTE TM1 1CR 20 MHz BW) 10.56 dB peak-to-rms ratio (one-carrier W-CDMA) 6.2 dB peak-to-rms ratio (64 QAM) CW input, TA = +25C, VTADJ = 0.45 V Calibration at -60, -45 and 0 dBm Calibration at -60, -45 and 0 dBm Deviation from output at 25C -40C < TA < +85C; PIN = 0 dBm -40C < TA < +85C; PIN = -45 dBm -40C < TA < +125C; PIN = 0 dBm -40C < TA < +125C; PIN = -45 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm -45 dBm < PIN < 0 dBm; calibration at -45 dBm and 0 dBm CW input, TA = +25C, VTADJ = 0.5 V Calibration at -60 dBm, -40 dBm, and 0 dBm Calibration at -60 dBm, -40 dBm, and 0 dBm
-0.1 -0.05 -0.05 64 3 -61 -0.1/0 -0.3/+0.3 -0.1/0 -0.3/+0.4 52.6 -62.6
dB dB dB dB dBm dBm dB dB dB dB mV/dB dBm
1.9 GHz 1.0 dB Dynamic Range Maximum Input Level, 1.0 dB Minimum Input Level, 1.0 dB Deviation vs. Temperature
Logarithmic Slope Logarithmic Intercept 2.14 GHz 1.0 dB Dynamic Range Maximum Input Level, 1.0 dB Minimum Input Level, 1.0 dB Deviation vs. Temperature
65 3 -62 -0.1/0 -0.3/+0.3 -0.1/0 -0.3/+0.4 52.4 -62.9 -0.1 -0.1 -0.1 -0.07 65 5 -60 0.4/0 +0.5/-0.6 0.6/0 +0.7/-0.6 51.0 -62.1
dB dBm dBm dB dB dB dB mV/dB dBm dB dB dB dB dB dBm dBm dB dB dB dB mV/dB dBm
Logarithmic Slope Logarithmic Intercept Deviation from CW Response
2.6 GHz 1.0 dB Dynamic Range Maximum Input Level, 1.0 dB Minimum Input Level, 1.0 dB Deviation vs. Temperature
Logarithmic Slope Logarithmic Intercept 3.5 GHz 1.0 dB Dynamic Range Maximum Input Level, 1.0 dB Minimum Input Level, 1.0 dB
57 8 -49
dB dBm dBm
Rev. 0 | Page 4 of 28
ADL5902
Parameter Test Conditions Min Typ Max Unit
Deviation vs. Temperature
Logarithmic Slope Logarithmic Intercept 5.8 GHz 1.0 dB Dynamic Range Maximum Input Level, 1.0 dB Minimum Input Level, 1.0 dB Deviation vs. Temperature
Deviation from output at 25C -40C < TA < +85C; PIN = 0 dBm -40C < TA < +85C; PIN = -40 dBm -40C < TA < +125C; PIN = 0 dBm -40C < TA < +125C; PIN = -40 dBm -40 dBm < PIN < 0 dBm; calibration at -30 dBm and 0 dBm -40 dBm < PIN < 0 dBm; calibration at -30 dBm and 0 dBm CW input, TA = +25C, VTADJ = 0.95 V Calibration at -50 dBm, -30 dBm, and 0 dBm Calibration at -50 dBm, -30 dBm, and 0 dBm Deviation from output at 25C -40C < TA < +85C; PIN = 0 dBm -40C < TA < +85C; PIN = -30 dBm -40C < TA < +125C; PIN = 0 dBm -40C < TA < +125C; PIN = -30 dBm -30 dBm < PIN < 0 dBm; calibration at -30 dBm and 0 dBm -30 dBm < PIN < 0 dBm; calibration at -30 dBm and 0 dBm VOUT (Pin 6) Swing range minimum, RL 500 to ground Swing range maximum, RL 500 to ground ILOAD = 8 mA, source/sink RFIN = 2.14 GHz, -20 dBm, fNOISE = 100 kHz, CLPF = 220 pF Transition from no input to 1 dB settling at PIN = -10 dBm, CLPF = 220 pF Transition from -10 dBm to off (1 dB of final value), CLPF = 220 pF VSET (Pin 7) Log conformance error 1 dB, minimum 2.14 GHz Log conformance error 1 dB, maximum 2.14 GHz f = 2.14 GHz f = 2.14 GHz Pin TADJ/PWDN (Pin 1) 0 VTADJ = 0.4 V VTADJ = 0.4 V VREF (Pin 11) PIN = -55 dBm 25C TA 125C -15C TA +25C -40C TA -15C 25C TA 125C -40C TA < +25C TA = 25C, ILOAD = 2 mA
0.2/0 -0.2/+0.4 +0.2/-0.3 -0.2/+0.4 49.6 -63.1
dB dB dB dB mV/dB dBm
61 9 -52 -0.8/0 -1.3/+0.1 -1.6/0 -1.3/+0.1 42.7 -54.1
dB dBm dBm dB dB dB dB mV/dB dBm
Logarithmic Slope Logarithmic Intercept OUTPUT INTERFACE Output Swing, Controller Mode Current Source/Sink Capability Voltage Regulation Output Noise Rise Time Fall Time SETPOINT INPUT Voltage Range Input Resistance Logarithmic Scale Factor Logarithmic Intercept TEMPERATURE COMPENSATION Input Voltage Range Input Bias Current Input Resistance VOLTAGE REFERENCE Output Voltage Temperature Sensitivity
0.03 4.8 10/10 +0.2/-0.2 25 3 25
V V mA % nV/Hz s s
3.5 0.23 72 52.4 -62.9 VS 2 200 2.3 -0.16 0.045 -0.04 4/0.05 3/0.05 -0.4
V V k mV/dB dBm V A k V mV/C mV/C mV/C mA mA %
Short-Circuit Current Source/ Sink Capability Voltage Regulation
Rev. 0 | Page 5 of 28
ADL5902
Parameter Test Conditions Min Typ Max Unit
TEMPERATURE REFERENCE Output Voltage Temperature Coefficient Short-Circuit Current Source/ Sink Capability Voltage Regulation RMS TARGET INTERFACE Input Voltage Range Input Bias Current Input Resistance POWER-DOWN INTERFACE Voltage Level to Enable Voltage Level to Disable Input Current
TEMP (Pin 8) TA = 25C, RL 10 k -40C TA +125C, RL 10 k 25C TA 125C -40C TA < +25C TA = 25C, ILOAD = 1 mA VTGT (Pin 12) 0.2 VTGT = 0.8 V Pin TADJ/PWDN (Pin 1) VPWDN decreasing VPWDN increasing VPWDN = 5 V VPWDN = 4.5 V VPWDN = 0 V VTADJ low to VOUT at 1 dB of final value, CLPA/B = 220 pF, PIN = 0 dBm VTADJ high to VOUT at 1 dB of final value, CLPA/B = 220 pF, PIN = 0 dBm VPOS (Pin 3, Pin 10) 4.5 TA = 25C, PIN < -60 dBm TA = 125C, PIN < -60 dBm VTADJ > VS - 0.1 V
1.4 4.9 4/0.05 3/0.05 -2.8 2.5 8 100 4 4.9 1 500 3 5 3
V mV/C mA mA % V A k V V A A A s s
Enable Time Disable Time POWER SUPPLY INTERFACE Supply Voltage Quiescent Current Power-Down Current
5 73 90 300
5.5
V mA mA A
Rev. 0 | Page 6 of 28
ADL5902 ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Supply Voltage, VPOS Input Average RF Power1 Equivalent Voltage, Sine Wave Input Internal Power Dissipation JC2 JB2 JA2 JT2 JB2 Maximum Junction Temperature Operating Temperature Range Storage Temperature Range Lead Temperature (Soldering, 60 sec)
1 2
ESD CAUTION
Rating 5.5 V 21 dBm 2.51 V p-p 550 mW 10.6C/W 35.3C/W 57.2C/W 1.0C/W 34C/W 150C -40C to +125C -65C to +150C 300C
This is for long durations. Excursions above this level, with durations much less than 1 second, are possible without damage. No airflow with the exposed pad soldered to a 4-layer JEDEC board.
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
Rev. 0 | Page 7 of 28
ADL5902 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
15 INLO 14 INHI 16 NC
PIN 1 INDICATOR
13 NC
TADJ/PWDN 1 NC 2 VPOS 3 COMM 4
12 VTGT 11 VREF 10 VPOS 9 COMM
ADL5902
TOP VIEW (Not to Scale)
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. 1 2 3, 10 4, 9, EPAD 5 6 7 Mnemonic TADJ/PWDN NC VPOS COMM CLPF VOUT VSET Description This is a dual function pin used for controlling the amount of nonlinear intercept temperature compensation at voltages <2.5 V and/or for shutting down the device at voltages >4 V. If the shutdown function is not used, this pin can be connected to the VREF pin through a voltage divider. See Figure 41 for an equivalent circuit. No Connect. Supply for the Device. Connect this pin to a 5 V power supply. Pin 3 and Pin 10 are not internally connected; therefore, both must connect to the source. System Common Connection. Connect these pins via low impedance to system common. The exposed paddle is also COMM and should have both a good thermal and good electrical connection to ground. Connection for RMS Averaging Capacitor. Connect a ground-referenced capacitor to this pin. A resistor can be connected in series with this capacitor to modify loop stability and response time. See Figure 43 for an equivalent circuit. Output. In measurement mode, this pin is connected to VSET. In controller mode, this pin can be used to drive a gain control element. See Figure 43 for an equivalent circuit. The voltage applied to this pin sets the decibel value of the required RF input voltage that results in zero current flow in the loop integrating capacitor pin, CLPF. This pin controls the variable gain amplifier (VGA) gain such that a 50 mV change in VSET changes the gain by approximately 1 dB. See Figure 42 for an equivalent circuit. Temperature Sensor Output of 1.4 V at 25C with a Coefficient of 5 mV/C. See Figure 38 for an equivalent circuit. General-Purpose Reference Voltage Output of 2.3 V at 25C. See Figure 39 for an equivalent circuit. The voltage applied to this pin determines the target power at the input of the RF squaring circuit. The intercept voltage is proportional to the voltage applied to this pin. The use of a lower target voltage increases the crest factor capacity; however, this may affect the system loop response. See Figure 44 for an equivalent circuit. No Connect. RF Input. The RF input signal is normally ac-coupled to this pin through a coupling capacitor. See Figure 37 for an equivalent circuit. RF Input Common. This pin is normally ac-coupled to ground through a coupling capacitor. See Figure 37 for an equivalent circuit. No Connect.
8 11 12 13 14 15 16
TEMP VREF VTGT NC INHI INLO NC
Rev. 0 | Page 8 of 28
08218-002
NOTES 1. NC = NO CONNECT. 2. THE EXPOSED PADDLE IS COMM AND SHOULD HAVE BOTH A GOOD THERMAL AND GOOD ELECTRICAL CONNECTION TO GROUND.
VOUT 6
TEMP 8
CLPF 5
VSET 7
ADL5902 TYPICAL PERFORMANCE CHARACTERISTICS
VS = 5 V, ZO = 50 , single-ended input drive, VOUT connected to VSET, VTGT = 0.8 V, CLPF = 0.1 F, TA = +25C (black), -40C (blue), +85C (red), +125C (orange) where appropriate. Error referred to the best fit line (linear regression) from - 10 dBm to - 40 dBm, unless otherwise indicated. Input RF signal is a sine wave (CW), unless otherwise indicated.
6.0 5.5 5.0 4.5
TADJ = 0.5V CALIBRATION AT 0dBm, -45dBm, AND -60dBm 6 5 4
6.0 5.5 5.0 4.5
VTADJ = 0.5V REPRESENTS 55 DEVICES FROM 2 LOTS
6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6
08218-006
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
3 2
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60
-50 -40 -30 -20 -10 0 10
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
1 0 -1 -2 -3 -4 -5
ERROR (dB)
08218-003
-6 PIN (dBm)
0 -60
-50
-40
-30
-20
-10
0
10
PIN (dBm)
Figure 3. Typical VOUT and Log Conformance Error with Respect to 25C Ideal Line over Temperature vs. Input Amplitude at 100 MHz, CW
6.0 5.5 5.0 4.5
TADJ = 0.4V CALIBRATION AT 0dBm, -45dBm, AND -60dBm 6 5 4
Figure 6. Distribution of Error with Respect to 25C over Temperature vs. Input Amplitude, CW, Frequency = 100 MHz
6.0 5.5 5.0 4.5 VTADJ = 0.4V REPRESENTS 55 DEVICES FROM 2 LOTS
6 5 4 3 2 1 0 -1 -2 -3 -4 -5
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
3 2
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60
-50 -40 -30 -20 -10 0 10
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
1 0 -1 -2 -3 -4 -5
ERROR (dB)
08218-004
-50
-40
-30
-20
-10
0
10
PIN (dBm)
PIN (dBm)
Figure 4. Typical VOUT and Log Conformance Error with Respect to 25C Ideal Line over Temperature vs. Input Amplitude at 700 MHz, CW
6.0 5.5 5.0 4.5
TADJ = 0.4V CALIBRATION AT 0dBm, -45dBm, AND -60dBm
Figure 7. Distribution of Error with Respect to 25C over Temperature vs. Input Amplitude, CW, Frequency = 700 MHz
6.0 5.5 5.0 4.5 VTADJ = 0.4V REPRESENTS 55 DEVICES FROM 2 LOTS
6 5 4
6 5 4 3 2 1 0 -1 -2 -3 -4 -5
08218-008
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
3 2
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60
-50 -40 -30 -20 -10 0
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
1 0 -1 -2 -3 -4 -5
08218-005
ERROR (dB)
-6 10
0 -60
-50
-40
-30
-20
-10
0
-6 10
PIN (dBm)
PIN (dBm)
Figure 5. Typical VOUT and Log Conformance Error with Respect to 25C Ideal Line over Temperature vs. Input Amplitude at 900 MHz, CW
Figure 8. Distribution of Error with Respect to 25C over Temperature vs. Input Amplitude, CW, Frequency = 900 MHz
Rev. 0 | Page 9 of 28
ERROR (dB)
08218-007
-6
0 -60
-6
ERROR (dB)
ERROR (dB)
ADL5902
6.0 5.5 5.0 4.5
TADJ = 0.4V CALIBRATION AT 0dBm, -45dBm, AND -60dBm
6 5 4
6.0 5.5 5.0 4.5
VTADJ = 0.4V REPRESENTS 55 DEVICES FROM 2 LOTS
6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6
08218-012
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
3 2
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60 -50 -40 -30 -20 -10 0
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
1 0 -1 -2 -3 -4 -5
ERROR (dB)
08218-009
-6 10
0 -60
-50
-40
-30
-20
-10
0
10
PIN (dBm)
PIN (dBm)
Figure 9. Typical VOUT and Log Conformance Error with Respect to 25C Ideal Line over Temperature vs. Input Amplitude at 1.9 GHz, CW
6.0 5.5 5.0 4.5
TADJ = 0.4V CALIBRATION AT 0dBm, -45dBm, AND -60dBm
Figure 12. Distribution of Error with Respect to 25C over Temperature vs. Input Amplitude, CW, Frequency = 1.9 GHz
6.0 5.5 5.0 4.5 VTADJ = 0.4V REPRESENTS 55 DEVICES FROM 2 LOTS
6 5 4
6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
3 2
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60 -50 -40 -30 -20 -10 0
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
1 0 -1 -2 -3 -4 -5
ERROR (dB)
08218-010
-50
-40
-30
-20
-10
0
10
PIN (dBm)
PIN (dBm)
Figure 10. Typical VOUT and Log Conformance Error with Respect to 25C Ideal Line over Temperature vs. Input Amplitude at 2.14 GHz, CW
6.0 5.5 5.0 4.5
TADJ = 0.45V CALIBRATION AT 0dBm, -45dBm, AND -60dBm
Figure 13. Distribution of Error with Respect to 25C over Temperature vs. Input Amplitude, CW, Frequency = 2.14 GHz
6.0 5.5 5.0 4.5 VTADJ = 0.45V REPRESENTS 55 DEVICES FROM 2 LOTS
6 5 4 3 2
6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6
08218-014
OUTPUT VOLTAGE (V)
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60 -50 -40 -30 -20 -10 0
OUTPUT VOLTAGE (V)
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
ERROR (dB)
1 0 -1 -2 -3 -4 -5
08218-011
-6 10
0 -60
-50
-40
-30
-20
-10
0
10
PIN (dBm)
PIN (dBm)
Figure 11. Typical VOUT and Log Conformance Error with Respect to 25C Ideal Line over Temperature vs. Input Amplitude at 2.6 GHz, CW
Figure 14. Distribution of Error with Respect to 25C over Temperature vs. Input Amplitude, CW, Frequency = 2.6 GHz
Rev. 0 | Page 10 of 28
ERROR (dB)
08218-013
-6 10
0 -60
ERROR (dB)
ERROR (dB)
ADL5902
6.0 5.5 5.0 4.5 TADJ = 0.5V CALIBRATION AT 0dBm, -40dBm, AND -60dBm 6 5 4 3 2
6.0 5.5 5.0 4.5 VTADJ = 0.5V REPRESENTS 55 DEVICES FROM 2 LOTS
6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6
08218-018
OUTPUT VOLTAGE (V)
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60 -50 -40 -30 -20 -10 0
OUTPUT VOLTAGE (V)
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60
-50 -40 -30 -20 -10 0 10
ERROR (dB)
1 0 -1 -2 -3 -4 -5 -6 10
PIN (dBm)
08218-115
PIN (dBm)
Figure 15. Typical VOUT and Log Conformance Error with Respect to 25C Ideal Line over Temperature vs. Input Amplitude at 3.5 GHz, CW
3.0
TADJ = 0.95V CALIBRATION AT 0dBm, -30dBm, AND -50dBm
Figure 18. Distribution of Error with Respect to 25C over Temperature vs. Input Amplitude, CW, Frequency = 3.5 GHz
3.0 VTADJ = 0.95V REPRESENTS 55 DEVICES FROM 2 LOTS
6 5 4 3
6 5 4 3
2.5
2.5
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
2.0
2
2.0
2 1
1
1.5
ERROR (dB)
0 -1
1.5
0 -1
1.0
-2 -3
1.0
-2 -3
0.5
-4 -5
08218-016
0.5
-4 -5
08218-019
08218-020
0 -60
-50
-40
-30
-20
-10
0
-6 10
0 -60
-6
-50 -40 -30 -20 -10 0 10
PIN (dBm)
PIN (dBm)
Figure 16. Typical VOUT and Log Conformance Error with Respect to 25C Ideal Line over Temperature vs. Input Amplitude at 5.8 GHz, CW
350 300 250 REPRESENTS 1900 PARTS FROM 3 LOTS
Figure 19. Distribution of Error with Respect to 25C over Temperature vs. Input Amplitude, CW, Frequency = 5.8 GHz
350 REPRESENTS 1900 PARTS FROM 3 LOTS 300 250 200 150 100 50 0 0.20
COUNT
200 150 100 50 0 2.65
08218-017
2.70
2.75
2.80
2.85 VOUT (V)
2.90
2.95
3.00
3.05
COUNT
0.25
0.30
0.35 VOUT (V)
0.40
0.45
0.50
Figure 17. Distribution of VOUT, PIN = -10 dBm, 900 MHz
Figure 20. Distribution of VOUT, PIN = -60 dBm, 900 MHz
Rev. 0 | Page 11 of 28
ERROR (dB)
ERROR (dB)
ADL5902
6.0 5.5 5.0 4.5
VOUT CW PEP = 0dB VOUT QPSK PEP = 2.76 VOUT 16 QAM PEP = 5.13 VOUT CDMA2000 PEP = 11.02 ERROR CW ERROR QPSK ERROR 16 QAM ERROR CDMA2000
6 5 4 3
6.0 5.5 5.0 4.5
VOUT CW PEP = 0dB VOUT 64 QAM PEP = 6.2dB VOUT 1CR W-CDMA PEP = 10.56dB VOUT 4CR W-CDMA VOUT LTE TM1 1CR 20MHz PEP = 11.58dB ERROR CW ERROR 64 QAM ERROR 1CR W-CDMA ERROR 4CR W-CDMA ERROR LTE TM1 1CR 20MHz
6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6
PIN (dBm)
08218-124
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -60
2
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
1 0 -1 -2 -3 -4 -5
ERROR (dB)
08218-121
-6
-50 -40 -30 -20 -10 0 10
0 -60
-50
-40
-30
-20
-10
0
10
PIN (dBm)
Figure 21. Error from CW Linear Reference vs. Signal Modulation, Frequency = 900 MHz, CLPF = 0.1F, Three-Point Calibration at 0 dBm, -45 dBm, and -60 dBm
6 RF ENVELOPE 0dBm -10dBm -20dBm -30dBm -40dBm
Figure 24. Error from CW Linear Reference vs. Signal Modulation, Frequency = 2.14 GHz, CLPF = 0.1 F, Three-Point Calibration at -10 dBm, -45 dBm, and -60 dBm
6 RF ENVELOPE 0dBm -10dBm 5
OUTPUT VOLTAGE (V)
-20dBm -30dBm -40dBm
5
OUTPUT VOLTAGE (V)
4
4
3
3
2
2
1
1
08218-027
-1
0
1
2
3
4
5
6
7
8
9
0
4
8
12
16
20
24
28
32
36
TIME (s)
TIME (s)
Figure 22. Output Response to RF Burst Input, Carrier Frequency 2.14 GHz, CLPF = 220 pF, Rising Edge
6 RF ENVELOPE 0dBm -10dBm -20dBm -30dBm -40dBm
Figure 25. Output Response to RF Burst Input, Carrier Frequency 2.14 GHz, CLPF = 220 pF, Falling Edge
6 RF ENVELOPE 0dBm -10dBm 5 -20dBm -30dBm -40dBm
5
OUTPUT VOLTAGE (V)
4
OUTPUT VOLTAGE (V)
4
3
3
2
2
1
1
08218-028
0
200
400
600
800
1000 1200 1400 1600 1800
-2000
0
2000
4000
6000
8000 10,000 12,000 14,000 16,000 18,000
TIME (s)
TIME (s)
Figure 23. Output Response to RF Burst Input, Carrier Frequency 2.14 GHz, CLPF = 0.1 F, Rising Edge
Figure 26. Output Response to RF Burst Input, Carrier Frequency 2.14 GHz, CLPF = 0.1 F, Falling Edge
Rev. 0 | Page 12 of 28
08218-031
0 -200
0
08218-030
0
0 -4
ERROR (dB)
ADL5902
REPRESENTS 1900 PARTS FROM 3 LOTS
2.5 2.3 2.1 1.9
2.5 2.0 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 -2.0
-35 -15 5 25 45 65 85 105 125
08218-036 08218-038
08218-035
400
300
VTEMP (V)
COUNT
1.7 1.5 1.3 1.1
200
100
0.9 0.7
08218-033
0 1.29
1.32
1.35
1.38
1.41
1.44
1.47
1.50
0.5 -55
-2.5
TEMPERATURE (C)
VTEMP VOLTAGE (V)
Figure 27. Distribution of VTEMP Voltage at 25C, No RF Input
Figure 30. VTEMP and Linearity Error with Respect to Straight Line vs. Temperature for Typical Device
0.2
REPRESENTS 1900 PARTS FROM 3 LOTS 400
0
CHANGE IN VREF (mV)
300
-0.2
COUNT
-0.4
200
-0.6
100
-0.8
08218-034
0 2.19
2.22
2.25
2.28
2.31
2.34
2.37
2.40
2.43
-1.0 -50
-40
-30
-20
-10
0
10
20
VREF BIAS VOLTAGE (V)
PIN (dBm)
Figure 28. Distribution of VREF Voltage at 25C, No RF Input
40
Figure 31. Change in VREF vs. Input Amplitude with Respect to -40 dBm, 25C, Typical Device
100
30 20
CHANGE IN VREF (mV)
10 0 -10 -20 -30 -40 -55
SUPPLY CURRENT (mA)
10
1
VPWDN DECREASING VPWDN INCREASING
-35
-15
5
25
45
65
85
105
125
08218-037
TEMPERATURE (C)
0.1 4.0
4.1
4.2
4.3
4.4
4.5
4.6
4.7
4.8
4.9
5.0
VPWDN (V)
Figure 29. Change in VREF vs. Temperature with Respect to 25C, RF Input = -40 dBm, Typical Device
Figure 32. Supply Current vs. VPWDN
Rev. 0 | Page 13 of 28
ERROR (C)
ADL5902
7 6
TADJ/PWDN PULSE 200 180
NOISE SPECTRAL DENSITY (nV/Hz)
160 140 120 100 80 60 40 20
OUTPUT VOLTAGE (V)
5
4 3
0dBm -10dBm
2 1 0 -4
-20dBm -30dBm -40dBm
08218-032
0
4
8
12
16
20
24
28
32
1k
10k
100k
1M
10M
TIME (s)
FREQUENCY (Hz)
Figure 33. Output Response Using Power-Down Mode for Various RF Input Levels Carrier Frequency 2.14 GHz, CLPF = 220 pF
3.5 3.0 2.5 -10dBm 2.0 1.5 1.0 0.5 0 0 1 2 3 4 5 6 7 8 9 FREQUENCY (GHz) -30dBm
Figure 35. Noise Spectral Density of VOUT, RF Input = -20 dBm, All CLPF Values
VOUT (V)
Figure 34. Typical VOUT vs. Frequency for Two RF Input Amplitudes, 50 MHz to 9 GHz
08218-026
Rev. 0 | Page 14 of 28
08218-039
0 100
ADL5902 THEORY OF OPERATION
The ADL5902 is a 50 MHz to 9 GHz true rms responding detector with a 65 dB measurement range at 2.14 GHz and a greater than 56 dB measurement range at frequencies up to 6 GHz. It incorporates a modified AD8362 architecture that increases the frequency range and improves measurement accuracy at high frequencies. Transfer function peak-to-peak ripple has been reduced to <0.1 dB over the entire dynamic range. Temperature stability of the rms output measurements provides <0.3 dB error, typically, over the specified temperature range of -40C to 125C through proprietary techniques. The device accurately measures waveforms that have a high peak-torms ratio (crest factor). The ADL5902 consists of a high performance AGC loop. As shown in Figure 36, the AGC loop comprises a wide bandwidth variable gain amplifier (VGA), square law detectors, an amplitude target circuit, and an output driver. For a more detailed description of the functional blocks, see the AD8362 data sheet. The nomenclature used in this data sheet to distinguish between a pin name and the signal on that pin is as follows: * * The pin name is all uppercase, for example, VPOS, COMM, and VOUT. The signal name or a value associated with that pin is the pin mnemonic with a partial subscript, for example, CLPF and VOUT. VGNS is a scaling voltage that defines the gain slope (the decibel change per voltage). The gain decreases with increasing VSET. The VGA output is VSIG = GSET x RFIN = GO x RFIN e
- (V SET / VGNS )
(2)
where RFIN is the ac voltage applied to the input terminals of the ADL5902. The output of the VGA, VSIG, is applied to a wideband square law detector. The detector provides the true rms response of the RF input signal, independent of waveform. The detector output, ISQR, is a fluctuating current with positive mean value. The difference between ISQR and an internally generated current, ITGT, is integrated by the parallel combination of CF and the external capacitor attached to the CLPF pin at the summing node. CF is an on-chip 26 pF filter capacitor, and CLPF, the external capacitance connected to the CLPF pin, can be used to arbitrarily increase the averaging time while trading off with the response time. When the AGC loop is at equilibrium Mean(ISQR) = ITGT This equilibrium occurs only when Mean(VSIG2) = VTGT2 (4) where VTGT is the voltage presented at the VTGT pin. This pin can conveniently be connected to the VREF pin through a voltage divider to establish a target rms voltage, VATG, of ~40 mV rms when VTGT = 0.8 V. Because the square law detectors are electrically identical and well matched, process and temperature dependent variations are effectively cancelled. (3)
SQUARE LAW DETECTOR AND AMPLITUDE TARGET
The VGA gain has the form GSET = GO e
- (V SET / VGNS )
(1)
where: GO is the basic fixed gain.
VPOS CH (INTERNAL) INHI VGA INLO GSET VSET VSIG
X2
SUMMING NODE ISQR ITGT
VATG = X2
VTGT 20 VTGT
CLPF CLPF (EXTERNAL) CF (INTERNAL) VOUT COMM TEMPERATURE COMPENSATION AND BIAS TEMPERATURE SENSOR BAND GAP REFERENCE TADJ/PWDN
TEMP (1.4V)
08218-040
VREF (2.3V)
Figure 36. Simplified Architecture Details
Rev. 0 | Page 15 of 28
ADL5902
When forcing the previous identity by varying the VGA setpoint, it is apparent that RMS(VSIG) = (Mean(VSIG2)) = (VATG2) = VATG Substituting the value of VSIG from Equation 2 results in RMS(G0 x RFIN e
(V SET / VGNS )
(5)
) = VATG
(6)
When connected as a measurement device, VSET = VOUT. Solving for VOUT as a function of RFIN, VOUT = VSLOPE x log10(RMS(RFIN)/VZ) where: VSLOPE is 1.06 V/decade (or 53 mV/dB) at 2.14 GHz. VZ is the intercept voltage. When RMS(RFIN) = VZ, this implies that VOUT = 0 V because log10(1) = 0. This makes the intercept the input that forces VOUT = 0 V if the ADL5902 had no sensitivity limit. The PINTERCEPT (in decibels relative to 1 milliwatt, that is, dBm) corresponding to Vz (in volts) in ADL5902 is given by the following equation: PINTERCEPT = -(VPEDISTAL/VSLOPE) + PMINDET (8) where VPEDISTAL is the VSET interface's pedestal voltage, and PMINDET is the minimum detectable signal in decibels relative to 1 milliwatt, given by the following expression: PMINDET = dBm (VATG) - GO (9) where dBm(VATG) is the equivalent power in decibels relative to 1 milliwatt corresponding to a given VTGT. Combining Equation 8 and Equation 9 results in PINTERCEPT = -(VPEDISTAL/VSLOPE) + dBm (VATG) - GO (10) (7)
half the supply voltage on each pin is established internally. Either the INHI or INLO pin can be used as the single-ended RF input pin. Signal coupling capacitors must be connected from the input signal to the INHI and INLO pins. A single external 60.4 resistor to ground from the desired input creates an equivalent 50 impedance over a broad section of the operating frequency range. The other input pin should be RF ac-coupled to common (ground). The input signal high-pass corner formed by the input coupling capacitor's internal and external resistances is fHIGHPASS = 1/(2 x x 50 x C) (11) where C is the capacitance in farads and fHIGHPASS is in hertz. The input coupling capacitors must be large enough in value to pass the input signal frequency of interest and determine the low end of the frequency response. INHI and INLO can also be driven differentially using a balun.
VBIAS
VPOS ESD 2k INHI LOAD 2k INLO ESD
ESD ESD ESD ESD ESD
ESD COMM
ESD
ESD
ESD
ESD
ESD
For the ADL5902, VPEDISTAL is approximately 0.275 V and VATG is given by VTGT/20. GO is 45 dB below approximately 4 GHz and then decreases at higher frequencies. VTGT = 0.8 V; therefore, VATG = 40 mV and dBm (VATG) = 10 log10((40 mV)2/50 )/1 mW) -14.9 dBm At 2.14 GHz, VSLOPE 53 mV/dB and GO at 2.14 GHz = 45 dB. This results in a PINTERCEPT -65 dBm. This differs slightly from the value in Table 1 due to the choice of calibration points and the slight nonideality of the response. In most applications, the AGC loop is closed through the setpoint interface and the VSET pin. In measurement mode, VOUT is directly connected to VSET (see the Measurement Mode Basic Connections section for more information). In controller mode, a control voltage is applied to VSET, and the VOUT pin typically drives the control input of an amplification or attenuation system. In this case, the voltage at the VSET pin forces a signal amplitude at the RF inputs of the ADL5902 that balances the system through feedback.
Figure 37. RF Inputs
Extensive ESD protection is employed on the RF inputs, and this protection limits the maximum possible input to the ADL5902.
SMALL SIGNAL LOOP RESPONSE
The ADL5902 uses a VGA in a loop to force a squared RF signal to be equal to a squared dc voltage. This nonlinear loop can be simplified and solved for a small signal loop response. The lowpass corner pole is given by FreqLP 1.83 x ITGT/(CLPF) where: ITGT is in amperes. CLPF is in farads. FreqLP is in hertz. ITGT is derived from VTGT; however, ITGT is a squared value of VTGT multiplied by a transresistance, namely ITGT = gm x VTGT2 (13) gm is approximately 18.9 s; therefore, with VTGT equal to the typically recommended 0.8 V, ITGT is approximately 12 A. The value of this current varies with temperature; therefore, the small signal pole varies with temperature. However, because the RF squaring circuit and dc squaring circuit track with temperature, (12)
RF INPUT INTERFACE
Figure 37 shows the RF input connections within the ADL5902. The input impedance is set primarily by an internal 2 k resistor connected between INHI and INLO. A dc level of approximately
Rev. 0 | Page 16 of 28
08218-041
ADL5902
there is no temperature variation contribution to the absolute value of VOUT. For CW signals, FreqLP 67.7 x 10-6/(CLPF) (14) However, signals with large crest factors include low pseudorandom frequency content that must be either filtered out or sampled and averaged out (see the Choosing a Value for CLPF section for more information). Table 4. Recommended VTADJ for Selected Frequencies
Frequency 100 MHz 700 MHz 900 MHz 1.9 GHz 2.14 GHz 2.6 GHz 3.5 GHz 5.8 GHz VTADJ (V) 0.5 0.4 0.4 0.4 0.4 0.45 0.5 0.95 R9 in Figure 54 () 1430 1430 1430 1430 1430 1430 1430 1430 R12 in Figure 54 () 402 301 301 301 301 348 402 1007
TEMPERATURE SENSOR INTERFACE
The ADL5902 provides a temperature sensor output with a scaling factor of the output voltage of approximately 4.9 mV/C. The output is capable of sourcing 4 mA and sinking 50 A maximum at 25C. An external resistor can be connected from TEMP to COMM to provide additional current sink capability. The typical output voltage at 25C is approximately 1.4 V.
VPOS INTERNAL VPAT TEMP 12k 4k
08218-042
The values in Table 4 were chosen to give the best drift performance at the high end of the usable dynamic range over the -40C to +85C temperature range. There is often a trade off in setting values, and optimizing for one area of the dynamic range may mean less than optimal drift performance at other input amplitudes. Compensating the device for temperature drift using TADJ allows for great flexibility. If the user requires minimum temperature drift at a given input power, a subset of the dynamic range, or even over a different temperature range than shown in this data sheet, the VTADJ can be swept while monitoring VOUT over the temperature at the frequency and amplitude of interest. The optimal VTADJ to achieve minimum temperature drift at a given power and frequency is the value of VTADJ where the output has minimum movement.
2.83 +125C 2.81 +105C +85C
VOUT (V)
COMM
Figure 38. TEMP Interface Simplified Schematic
VREF INTERFACE
The VREF pin provides an internally generated voltage reference for the user. The VREF voltage is a temperature stable 2.3 V reference that is capable of sourcing 4 mA and sinking 50 A maximum. An external resistor can be connected from VREF to COMM to provide additional current sink capability. The voltage on this pin can be used to drive the TADJ/PWDN and VTGT pins.
VPOS INTERNAL VOLTAGE VREF
2.79
+55C +25C 0C
2.77
-20C -40C
2.75
16k
08218-143
0.2
0.3
0.4
0.5
0.6
0.7
0.8
VTADJ (V)
COMM
Figure 40. Effect of VTADJ at Various Temperatures, 2.14 GHz, -10 dBm
Figure 39. VREF Interface Simplified Schematic
TEMPERATURE COMPENSATION INTERFACE
While the ADL5902 has a highly stable measurement output with respect to temperature using proprietary techniques, for optimal performance, the output temperature drift must be compensated for using the TADJ pin. The absolute value of compensation varies with frequency and VTGT. Table 4 shows the recommended voltages for VTADJ to maintain a temperature drift error of typically 0.5 dB or better over the intended temperature range (-40C < TA < +85C) when driven single-ended and VTGT = 0.8 V.
Varying VTADJ has only a very slight effect on VOUT at device temperatures near 25C; however, the compensation circuit has more and more effect as the temperature departs farther from 25C. The TADJ pin has a high input impedance and can be conveniently driven from an external source or from an attenuated value of VREF using a resistor divider. Table 4 gives suggested voltage divider values to generate the required voltage from VREF. The resistors are shown in the evaluation board schematic (see Figure 54). VREF does change slightly with temperature and also input RF amplitude; however, the amount of change is unlikely to result in a significant effect on the final temperature
Rev. 0 | Page 17 of 28
08218-044
2.73 0.1
ADL5902
stability of the RF measurement system. Typically, the temperature compensation circuit responds only to voltages between 0 and VS/2, or about 2.5 V when VS = 5 V. Figure 41 in the Power-Down Interface section shows a simplified schematic representation of the TADJ/PWDN interface. no load is approximately 58 MHz with a single-pole roll off of approximately -20 dB/decade. The output noise is approximately 25 nV/Hz at 100 kHz. The VOUT pin can source and sink up to 10 mA. There is also an internal load from VOUT to COMM of 2500 .
VPOS
POWER-DOWN INTERFACE
The quiescent and disabled currents for the ADL5902 at 25C are approximately 73 mA and 300 A, respectively. The dual function TADJ/PWDN pin is connected to the temperature compensation circuit as well as the power-down circuit. Typically, the temperature compensation circuit responds only to voltages between 0 and VS/2, or about 2.5 V when VS = 5 V. When the voltage on this pin is greater than VS - 0.1 V, the device is fully powered down. Figure 32 shows this characteristic as a function of VPWDN. Note that, because of the design of this section of the ADL5902, as VPWDN passes through a narrow range at ~4.5 V (or ~VS - 0.5 V), the TADJ/PWDN pin sinks approximately 500 A. The source used to disable the ADL5902 must have a sufficiently high current capability for this reason. Figure 33 shows the typical response times for various RF input levels. The output reaches within 0.1 dB of its steadystate value in approximately 5 s; however, the reference voltage is available to full accuracy in a much shorter time. This wake-up response varies depending on the input coupling and CLPF.
VPOS ESD SHUTDOWN POWER-UP CIRCUIT CIRCUIT TADJ/ PWDN 200 ESD 7k 200 7k VREF
ESD 2pF
CLPF
ESD 2k ESD
VOUT
COMM
Figure 43. VOUT Interface Simplified Schematic
VTGT INTERFACE
The target voltage can be set with an external source or by connecting the VREF pin (nominally 2.3 V) to the VTGT pin through a resistive voltage divider. With 0.8 V on the VTGT pin, the rms voltage that must be provided by the VGA to balance the AGC feedback loop is 0.8 V x 0.05 = 40 mV rms. Most of the characterization information in this data sheet was collected at VTGT = 0.8 V. Voltages higher and lower than this can be used; however, doing so increases or decreases the gain at the internal squaring cell, which results in a corresponding increase or decrease in intercept. This, in turn, affects the sensitivity and the usable measurement range, in addition to the sensitivity to different carrier modulation schemes. As VTGT decreases, the squaring circuits produce more noise; this becomes noticeable in the output response at low input signal amplitudes. As VTGT increases, measurement error due to modulation increases and temperature drift tends to decrease. The chosen VTGT value of 0.8 V represents a compromise between these characteristics.
VPOS ESD g x X2 VTGT 50k ITGT
200 ESD COMM
Figure 41. TADJ/PWDN Interface Simplified Schematic
VSET INTERFACE
The VSET interface has a high input impedance of 72 k. The voltage at VSET is converted to an internal current used to set the internal VGA gain. The VGA attenuation control is approximately 19 dB/V.
GAIN ADJUST VSET 54k
08218-076
INTERCEPT TEMPERATURE COMPENSATION
ESD 50k ESD 10k
08218-045 08218-048
500
18k 2.5k
08218-149
COMM
Figure 44. VTGT Interface
BASIS FOR ERROR CALCULATIONS
The slope and intercept used in the error plots are calculated using the coefficients of a linear regression performed on data collected in its central operating range. The error plots in the Typical Performance Characteristics section are shown in two formats: error from the ideal line and error with respect to the 25C output voltage. The error from the ideal line is the decibel difference in VOUT from the ideal straight-line fit of VOUT
ACOM
Figure 42. VSET Interface Simplified Schematic
OUTPUT INTERFACE
The ADL5902 incorporates rail-to-rail output drivers with pullup and pull-down capabilities. The closed-loop, - 3dB bandwidth from the input of the output amplifier to the output with
Rev. 0 | Page 18 of 28
ADL5902
calculated by the linear-regression fit over the linear range of the detector, typically at 25C. The error in decibels is calculated by Error (dB) = (VOUT - Slope x (PIN - PZ))/Slope (15) where PZ is the x-axis intercept expressed in decibels relative to 1 milliwatt (the input amplitude that would produce a 0 V output if such an output were possible). The error from the ideal line is not a measure of absolute accuracy because it is calculated using the slope and intercept of each device. However, it verifies the linearity and the effect of temperature and modulation on the response of the device. An example of this type of plot is Figure 3. The slope and intercept that form the ideal line are those at 25C with CW modulation. Figure 21 and Figure 24 show the error with various popular forms of modulation with respect to the ideal CW line. This method for calculating error is accurate, assuming that each device is calibrated at room temperature. In the second plot format, the VOUT voltage at a given input amplitude and temperature is subtracted from the corresponding VOUT at 25C and then divided by the 25C slope to obtain an error in decibels. This type of plot does not provide any information on the linear-in-dB performance of the device; it merely shows the decibel equivalent of the deviation of VOUT over temperature, given a calibration at 25C. When calculating error from any one particular calibration point, this error format is accurate. It is accurate over the full range shown on the plot assuming that enough calibration points are used. Figure 6 shows this plot type.
C3 0.1F 5V 5V
The error calculations for Figure 30 are similar to those for the VOUT plots. The slope and intercept of the VTEMP function vs. temperature are determined and applied as follows: Error (C) = (VTEMP - Slope x (Temp - TZ))/Slope (16) where: TZ is the x-axis intercept expressed in degrees Celsius (the temperature that would result in a VTEMP of 0 V if this were possible). Temp is the ambient temperature of the ADL5902 in degrees Celsius. Slope is, typically, 4.9 mV/C. VTEMP is the voltage at the TEMP pin at that temperature.
MEASUREMENT MODE BASIC CONNECTIONS
The ADL5902 requires a single supply of nominally 5 V. The supply is connected to the two VPOS supply pins. These pins should each be decoupled using the two capacitors with values equal or similar to those shown in Figure 45. These capacitors should be placed as close as possible to the VPOS pins. An external 60.4 resistor combines with the relatively high RF input impedance of the ADL5902 to provide a broadband 50 match. An ac coupling capacitor should be placed between this resistor and INHI. The INLO input should be ac-coupled to ground using the same value capacitor. Because the ADL5902 has a minimum input operating frequency of 50 MHz, 100 pF ac coupling capacitors can be used. The ADL5902 is placed in measurement mode by connecting VOUT to VSET. In measurement mode, the output voltage is proportional to the log of the rms input signal level.
C7 0.1F
C4 100pF
C5 100pF
VPOS
3
POS
10
ADL5902
C10 100pF RFIN R3 60.4 INHI INLO C12 100pF
TEMPERATURE SENSOR
8
TEMP VSET VOUT
7
14
X2
15
IDET
LINEAR-IN-dB VGA (NEGATIVE SLOPE)
2
X2
ITGT G=5
6
NC
VOUT
NC 16
BIAS AND POWERDOWN CONTROL
VREF 2.3V
5
CLPF
NC 13
26pF
1
11
12
9
4
TADJ/PWDN
VREF R9 (SEE TABLE 4)
VTGT R10 3.74k
COMM
COMM
C9 10F (SEE THE CHOOSING A VALUE FOR CLPF SECTION.)
Figure 45. Basic Connections for Operation in Measurement Mode
Rev. 0 | Page 19 of 28
08218-145
R12 (SEE TABLE 4)
R11 2k
ADL5902
SETTING VTADJ
As discussed in the Theory of Operation section, the output temperature drift must be compensated by applying a voltage to the TADJ pin. The compensating voltage varies with frequency. The voltage for the TADJ pin can be easily derived from a resistor divider connected to the VREF pin. Table 5 shows the recommended VTADJ for operation from -40C to +85C, along with resistor divider values. Resistor values are chosen so that they neither pull too much current from VREF (VREF short-circuit current is 4 mA) nor are so large that the TADJ pin's bias current of 3 A affects the resulting voltage at the TADJ pin. Table 5. Recommended VTADJ for Selected Frequencies
Frequency 100 MHz 700 MHz to 2.14 GHz 2.6 GHz 3.5 GHz 5.8 GHz VTADJ (V) 0.5 0.4 0.45 0.5 0.95 R9 () 1430 1430 1430 1430 1430 R12 () 402 301 348 402 1007
Figure 46 shows how output noise varies with CLPF when the ADL5902 is driven by a single-carrier W-CDMA signal (Test Model TM1-64, peak envelope power = 10.56 dB, bandwidth = 3.84 MHz). With a 10 F capacitor on CLPF, there is residual noise on VOUT of 4.4 mV p-p, which is less than 0.1 dB error (assuming a slope of approximately 53 mV/dB).
300
OUTPUT NOISE (mV p-p) 10% TO 90% RISE TIME (s) 90% TO 10% FALL TIME (s)
1M
250
100k
OUTPUT NOISE (mV p-p)
200
10k
150
1k
100
100
50
10
0
1 10 CLPF (nF) 100
SETTING VTGT
As discussed in the Theory of Operation section, setting the voltage on VTGT to 0.8 V represents a compromise between achieving excellent rms compliance and maximizing dynamic range. The voltage on VTGT can be derived from the VREF pin using a resistor divider as shown Figure 45. Like the resistors chosen to set the VTADJ voltage, the resistors setting VTGT should have reasonable values that do not pull too much current from VREF or cause bias current errors. Also, attention should be paid to the combined current that VREF must deliver to generate the VTADJ and VTGT voltages. This current should be kept well below the VREF short-circuit current of 4 mA.
Figure 46. Output Noise, Rise and Fall Times vs. CLPF Capacitance, SingleCarrier W-CDMA (TM1-64) at 2.14 GHz with PIN = 0 dBm
Figure 46 also shows how the response time is affected by the value of CLPF. To measure this, a RF burst at 2.14 GHz at -10 dBm was applied to the ADL5902. The 10% to 90% rise time and 90% to 10% fall time were then measured. It is notable that the fall time is much longer than the rise time. This can also be seen in the response time plots, Figure 22, Figure 23, Figure 25, and Figure 26. In applications where the response time is critical, a different approach to signal filtering can be taken. This is shown in Figure 47. The capacitor on the CLPF pin is set to the minimum value that ensures that a valid rms computation has been performed. The job of noise removal is then handed off to an RC filter on the VOUT pin. This approach ensures that there is enough averaging to ensure good rms compliance and does not burden the rms computation loop with extra filtering that will significantly slow down the response time. By finishing the filtering process using an RC filter after VOUT, faster fall times can be achieved with an equivalent amount of output noise. It should be noted that the RC filter can also be implemented in the digital domain after the analog-to-digital converter.
CHOOSING A VALUE FOR CLPF
CLPF provides the averaging function for the internal rms computation. Using the minimum value for CLPF allows the quickest response time to a pulsed waveform but leaves significant output noise on the output voltage signal. By the same token, a large filter cap reduces output noise but at the expense of response time. For non response-time critical applications, a relatively large capacitor can be placed on the CLPF pin. In Figure 45, a value of 10 F is used. For most signal modulation schemes, this value ensures excellent rms measurement compliance and low residual output noise. There is no maximum capacitance limit for CLPF.
Rev. 0 | Page 20 of 28
08218-146
1 1000
RISE/FALL TIME (s)
ADL5902
VPOS
3
POS
10
ADL5902
TEMPERATURE SENSOR
8
TEMP VSET
7
INHI
14
X2 INLO 15 LINEAR-IN-dB VGA (NEGATIVE SLOPE) NC
2
IDET
X2
ITGT G=5
6
RFILTER 2k VOUT VOUT CFILTER (SEE FIGURE 48.)
NC 16
BIAS AND POWERDOWN CONTROL
VREF 2.3V
5
NC 13
26pF
1
11
TADJ/PWDN
VREF
VTGT
COMM
COMM
Figure 47. Optimizing Setting Time and Residual Ripple
In Figure 47, CLPF is equal to 10 nF. This value was experimentally determined to be the minimum capacitance that ensures good rms compliance when the ADL5902 is driven by a 1 C W-CDMA signal (TM1-64). This test was carried out by starting out with a large capacitance value on the CLPF pin (for example, 10 F). The value of VOUT was noted for a fixed input power level (for example, -10 dBm). The value of CLPF was then progressively reduced (this can be done with press-down capacitors) until the value of VOUT started to deviate from its original value (this indicates that the accuracy of the rms computation is degrading and that CLPF is getting too small).
Figure 48 shows the resulting rise and fall times (signal is pulsed
For large values of CFILTER, the fall time is dramatically reduced compared to Figure 46. This comes at the expense of a moderate increase in rise time. As CFILTER is reduced, the fall time flattens out. This is because the fall time is now dominated by the 10 nF CLPF which is present throughout the measurement. Table 6 shows recommended minimum values of CLPF for popular modulation schemes, using just a single filter capacitor at the CLPF pin. Using lower capacitor values results in rms measurement errors. Output response time (10% to 90%) is also shown. If the output noise shown in Table 6 is unacceptably high, it can be reduced by * * * Increasing CLPF Adding an RC filter at VOUT, as shown in Figure 47 Implementing an averaging algorithm after the ADL5902's output voltage has been digitized by an ADC
between off and -10 dBm) with CLPF equal to 10 nF. A 2 k resistor is placed in series with the VOUT pin, and the capacitance from this resistor to ground (CFILTER in Figure 47) is varied up to 1 F.
300
RESIDUAL RIPPLE (V p-p) 10% TO 90% RISE TIME (s) 90% TO 10% FALL TIME (s)
1M
250
100k
RESIDUAL RIPPLE (mV p-p)
200
10k
150
1k
100
100
50
10
1
10
100
CFILTER (nF)
Figure 48. Residual Ripple, Rise and Fall Times Using an RC Low-Pass Filter at VOUT, PIN = 0 dBm at 2.14 GHz
Rev. 0 | Page 21 of 28
08218-148
0
1 1k
RISE/FALL TIME (s)
08218-147
12
9
4
CLPF C9 10nF (SEE TABLE 6 AND FIGURE 46.)
ADL5902
Table 6. Recommended Minimum CLPF Values for Various Modulation Schemes
Modulation/Standard W-CDMA, One-Carrier, TM1-64 W-CDMA Four-Carrier, TM1-64, TM1-32, TM1-16, TM1-8 LTE, TM1 1CR 20 MHz (2048 Subcarriers, QPSK Subcarrier Modulation) Peak-Envelope Power 10.56 dB 12.08 dB 11.58 dB Signal Bandwidth 3.84 MHz 18.84 MHz 20 MHz CLPF (min) 10 nF 5.6 nF 1000 pF Output Noise 95 mV p-p 164 mV p-p 452 mV p-p Rise/Fall Time (10% to 90%) 12/330 s 7/200 s 1.3/38 s
OUTPUT VOLTAGE SCALING
The output voltage range of the ADL5902 (nominally 0.3 V to 3.5 V) can be easily increased or decreased. There are a number of situations where adjustment of the output scaling makes sense. For example, if the ADL5902 is driving an analog-todigital converter (ADC) with a 0 V to 5 V input range, it makes sense to increase the detector's nominal maximum output voltage of 3.5 V so that it is closer to 5 V. This makes better use of the input range of the ADC and maximizes the resolution of the system in terms of bits/dB. If only a part of the ADL5902's RF input power range is being used (for example, -10 dBm to -60 dBm), it may make sense to increase the scaling so that this reduced input range fits into the ADL5902's available output swing of 0 V to 4.8 V. The output swing can also be reduced by simply adding a voltage divider on the output pin, as shown in Figure 49. Reducing the output scaling may be used when interfacing the ADL5902 to an ADC with a 0 V to 2.5 V input range. The output voltage swing can be increased using a technique that is analogous to setting the gain of an op amp in noninverting mode with the VSET pin being the equivalent of the inverting input of the op amp. Connecting VOUT to VSET results in the nominal 0 V to 3.5 V swing and a slope of approximately 53 mV/dB (this varies slightly with frequency). Figure 49 and Table 7 show the configurations for increasing the slope, along with recommended standard resistor values for particular input ranges and output swings.
Table 7. Output Voltage Range Scaling
Desired Input Range (dBm) 0 to -60 -10 to -50 0 to -60 -10 to -50 R1 () 665 1180 806 324 R2 () 2000 2000 2000 2000 New Slope (mV/dB) 72.1 86.3 38.3 46.2 Nominal Output Voltage Range (V) 0.195 to 4.52 1.096 to 4.55 0.103 to 2.49 0.587 to 2.43
Equation 17 is the general function that governs this.
V ' R1 = ( R2 || RIN ) O - 1 V O
where: VO is the nominal maximum output voltage (see Figure 6 through Figure 18). V'O is the new maximum output voltage (for example, up to 4.8 V). RIN is the VSET input resistance (72 k).
(17)
When choosing R1 and R2, attention must be paid to the current drive capability of the VOUT pin and the input resistance of the VSET pin. The choice of resistors should not result in excessive current draw out of VOUT. However, making R1 and R2 too large is also problematic. If the value of R2 is compatible with the input resistance of the VSET input (72 k), this input resistance, which will vary slightly from part to part, contributes to the resulting slope and output voltage. In general, the value of R2 should be at least ten times smaller than the input resistance of VSET. Values for R1 and R2 should, therefore, be in the 1 k to 5 k range. It is also important to take into account part-to-part and frequency variation in output swing along with the ADL5902 output stage's maximum output voltage of 4.8 V. The VOUT distribution is well characterized at major frequencies' bands in the Typical Performance Characteristics section (see Figure 6 through Figure 8, Figure 12 through Figure 14, Figure 18, and Figure 19). The resistor values in Table 7, which were calculated based on 900 MHz performance, are conservatively chosen so that there is no chance that the output voltages exceed the ADL5902 output swing or the input range of a 0 V to 2.5 V and 0 V to 5 V ADC. Because the output swing does not vary much with frequency (it does start to drop off above 3 GHz), these values work for multiple frequencies.
7
VSET
7
VSET R1
6
VOUT
R1 R2
6
VOUT R2
08218-049
Figure 49. Decreasing and Increasing Slope
Rev. 0 | Page 22 of 28
ADL5902
SYSTEM CALIBRATION AND ERROR CALCULATION
The measured transfer function of the ADL5902 at 2.14 GHz is shown in Figure 50, which contains plots of both output voltage vs. input amplitude (power) and calculated error vs. input level. As the input level varies from -62 dBm to +3 dBm, the output voltage varies from ~0.25 V to ~3.5 V.
6
VOUT ERROR 2-POINT CAL AT 0dBm, AND 40dBm ERROR 3-POINT CAL AT 0 dBm, -45dBm, AND 60dBm ERROR 4-POINT CAL AT 0dBm, -20dBm, -45dBm, AND -60dBm
error at the calibration points (in this case, -40 dBm and 0dBm) is equal to 0 by definition. The residual nonlinearity of the transfer function that is apparent in the two-point calibration error plot can be reduced by increasing the number of calibration points. Figure 50 shows the postcalibration error plots for three-point and four-point calibrations. With a multipoint calibration, the transfer function is segmented, with each segment having its own slope and intercept. Multiple known power levels are applied, and multiple voltages are measured. When the equipment is in operation, the measured voltage from the detector is first used to determine which of the stored slope and intercept calibration coefficients are to be used. Then the unknown power level is calculated by inserting the appropriate slope and intercept into Equation 21. Figure 51 shows the output voltage and error at 25C and over temperature when a four-point calibration is used (calibration points are 0 dBm, -20 dBm, -45 dBm, and -60 dBm). When choosing calibration points, there is no requirement for, or value, in equal spacing between the points. There is also no limit to the number of calibration points used. However, using more calibration points increases calibration time.
6
+85C VOUT +25C VOUT -40C VOUT +85C ERROR 4-POINT CAL +25C ERROR 4-POINT CAL AT 0dBm, -20dBm, -45dBm, AND -60dBm -40C ERROR 4-POINT CAL
6 5 4 3 2 1
5
4
3
0 -1
2
-2 -3
1
-4 -5
-60 -50 -40 -30 PIN (dBm) -20 -10 0 10
08218-050
0 -70
-6
Figure 50. 2.14 GHz Transfer Function, Using Various Calibration Techniques
Because slope and intercept vary from device to device, boardlevel calibration must be performed to achieve high accuracy. The equation for the idealized output voltage can be written as VOUT(IDEAL) = Slope x (PIN - Intercept) (18) where: Slope is the change in output voltage divided by the change in input power (dB). Intercept is the calculated input power level at which the output voltage is 0 V (note that Intercept is an extrapolated theoretical value not a measured value). In general, calibration is performed during equipment manufacture by applying two or more known signal levels to the input of the ADL5902 and measuring the corresponding output voltages. The calibration points are generally within the linearin-dB operating range of the device. With a two-point calibration, the slope and intercept are calculated as follows: Slope = (VOUT1 - VOUT2)/(PIN1 - PIN2) Intercept = PIN1 - (VOUT1/Slope) (19) (20)
ERROR (dB)
VOUT (V)
6 5 4 3 2 1
5
4
3
0 -1
2
-2 -3
1
-4 -5
-60 -50 -40 -30
PIN (dBm)
-20
-10
0
Figure 51. 2.14 GHz Transfer Function and Error at +25C, -40C, and +85C Using a Four-Point Calibration (0 dBm, -20 dBm, -45 dBm, -60 dBm)
The -40C and +85C error plots in Figure 51 are generated using the 25C calibration coefficients. This is consistent with equipment calibration in a mass production environment where calibration at just a single temperature is practical.
After the slope and intercept are calculated and stored in nonvolatile memory during equipment calibration, an equation can be used to calculate an unknown input power based on the output voltage of the detector. PIN (Unknown) = (VOUT1(MEASURED)/Slope) + Intercept The log conformance error is the difference between this straight line and the actual performance of the detector. Error (dB) = (VOUT(MEASURED) - VOUT(IDEAL))/Slope (22) Figure 50 includes a plot of this error when using a two-point calibration (calibration points are 0 dBm and -40 dBm). The (21)
HIGH FREQUENCY PERFORMANCE
The ADL5902 is specified to 6 GHz; however, operation is possible to as high as 9 GHz with sufficient dynamic range for many purposes. Figure 52 shows the typical VOUT response and conformance error at 7 GHz, 8 GHz, and 9 GHz.
Rev. 0 | Page 23 of 28
08218-051
0 -70
-6 10
ERROR (dB)
VOUT (V)
ADL5902
3.00 2.75 2.50 2.25
7GHz 8GHz 9GHz 6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6
08218-057
was driven in a single-ended configuration for most characterization, except where noted. Much of the data was taken using an Agilent E4438C signal source as a RF input stimulus. Several ADL5902 devices mounted on circuit boards constructed of Rodgers 3006 material were put into a test chamber simultaneously, and a Keithley S46 RF switching network connected the signal source to the appropriate device under test. The test chamber temperature was set to cycle over the appropriate temperature range. The signal source, switching, and chamber temperature were all controlled by a PC running Agilent VEE Pro. The subsequent response to stimulus was measured with a voltmeter and the results stored in a database for analysis later. In this way, multiple ADL5902 devices were characterized over amplitude, frequency, and temperature in a minimum amount of time. The RF stimulus amplitude was calibrated up to the circuit board that carries the ADL5902, and, thus, it does not account for the slight losses due to the connector on the circuit board that carries the ADL5902 nor for the loss of traces on the circuit board. For this reason, there is a small absolute amplitude error (generally <0.5 dB) not accounted for in the characterization data, but this is generally not important because the ADL5902's relative accuracy is unaffected.
OUTPUT VOLTAGE (V)
2.00 1.75 1.50 1.25 1.00 0.75 0.50 0.25 0 -50
-40 -30 -20 -10 0 10
PIN (dBm)
Figure 52. Typical VOUT and Log Conformance Error at 7 GHz, 8 GHz, and 9 GHz, 25C Only
LOW FREQUENCY PERFORMANCE
The lowest frequency of operation of the ADL5902 is approximately 50 MHz. This is the result of the circuit design and architecture of the ADL5902.
DESCRIPTION OF CHARACTERIZATION
The general hardware configuration used for most of the ADL5902 characterization is shown in Figure 53. The ADL5902
AGILENT E3631A DC POWER SUPPLIES
AGILENT 34980A SWITCH MATRIX/ DC METER
ERROR (dB)
ADL5902 CHARACTERIZ ATION BOARD - TEST SITE 1
AGILENT E8251A MICROWAVE SIGNAL GENERATOR
KEITHLEY S46 MICROWAVE SWITCH
ADL5902 CHARACTERIZ ATION BOARD - TEST SITE 2
RF
DC
DATA AND CONTROL
Figure 53. General Characterization Configuration
Rev. 0 | Page 24 of 28
08218-075
PERSONAL COMPUTER
ADL5902 CHARACTERIZ ATION BOARD - TEST SITE 3
ADL5902 EVALUATION BOARD SCHEMATICS AND ARTWORK
VTGT VREF VPOS2 C7 0.1F R7 0 R11 1k R10 1.87k VPOS R8 0 R14 0 C5 100pF
12
11
10
9
VTGT
VPOS
COMM
VREF
TEMP R2 OPEN TEMP 8 7 6 5 C9 0.1F R5 0 PADDLE AGND C4 100pF GND R6 0
VSET
C10 0.1F IN R3 60.4
C11 OPEN
13 14 15
R13 OPEN
VOUT
NC INHI INLO
ADL5902
DUT1
TADJ/PWDN
VSET VOUT CLPF
R1 0
R15 0 VOUTP
C6 OPEN
C12 0.1F
16
NC
COMM
VPOS
NC
1 TC2_PWDN
2
3
4
C8 OPEN
GND
R12 301
R9 1430 VPOSC VREF
R16 0
C13 0.1F
08218-059
VPOS1
Figure 54. Evaluation Board Schematic
Table 8. Evaluation Board Configuration Options
Component C6, C10, C11, C12, R3 Function/Notes RF input. The ADL5902 is generally driven single-ended. When driving INHI, populate C10 and C12 with an appropriate capacitor value for the frequency of operation and leave C6 and C11 open. When driving INLO, populate C6 and C11 with an appropriate capacitor value for the frequency of operation and leave C10 and C12 open. R3 is the input termination resistor and is chosen to give a 50 input impedance over a broad frequency range. VTGT interface. R10 and R11 are set up to provide 0.8 V to VTGT derived from VREF. If R10 and R11 are removed, an external voltage can be applied. Alternatively, R7 and R11 can be used to form a voltage divider for an external reference. Power supply decoupling. The nominal supply decoupling consists of a 100 pF filter capacitor placed close to the ADL5902, a 0 series resistor, and a 0.1 F capacitor placed close to the power supply input pin. Output interface. In measurement mode, a portion of the voltage at the VOUT pin is fed back to the VSET pin via R6. Using the voltage divider created by R2 and R6, the magnitude of the slope of VOUT is increased by reducing the portion of VOUT that is fed back to VSET. In controller mode, R6 must be open and R13 must have a 0 resistor. In this mode, the ADL5902 can control the gain of an external component. A setpoint voltage is applied to the VSET pin, the value of which corresponds to the desired RF input signal level applied to the ADL5902. Low-pass filter capacitors, CLPF. The low-pass filter capacitors reduce the noise on the output and affect the response time of the ADL5902. TADJ/PWDN. The TADJ/PWDN pin controls the amount of nonlinear intercept temperature compensation and/or shuts down the device. The evaluation board is configured with TADJ connected to VREF through a resistor divider (R9, R12). The paddle should be tied to both a thermal ground and an electrical ground. Default Value C6 = open, C10 = C12 = 0.1 F, C11 = open, R3 = 60.4 R7 = R8 = 0 , R10 = 1.87 k, R11 = 1k C4 = C5 = 100 pF, C7 = C13 = 0.1 F, R14 = R16 = 0 R1 = R6 = R15 = 0 , R2 = R13 = open
R7, R8, R10, R11 C4, C5, C7, C13, R14, R16 R1, R2, R6, R13, R15
C8, C9, R5 R9, R12 Paddle
C8 = open, C9 = 0.1 F, R5 = 0 , R9 = 1430 R12 = 301
Rev. 0 | Page 25 of 28
ADL5902
ASSEMBLY DRAWINGS
08218-060
Figure 55. Evaluation Board Layout, Top Side
Figure 56. Evaluation Board Layout, Bottom Side
Rev. 0 | Page 26 of 28
08218-061
ADL5902 OUTLINE DIMENSIONS
4.00 BSC SQ 0.60 MAX
13 16
0.60 MAX PIN 1 INDICATOR
1
PIN 1 INDICATOR
TOP VIEW
0.65 BSC 3.75 BSC SQ 0.50 0.40 0.30
12
EXPOSED PAD
(BOTTOM VIEW)
2.50 2.35 SQ 2.20
4
9 8 5
0.25 MIN 1.95 BSC
12 MAX 1.00 0.85 0.80
0.80 MAX 0.65 TYP 0.05 MAX 0.02 NOM
COMPLIANT TO JEDEC STANDARDS MO-220-VGGC
Figure 57. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ] 4 mm x 4 mm Body, Very Thin Quad (CP-16-10) Dimensions shown in millimeters
ORDERING GUIDE
Model1 ADL5902ACPZ-R7 ADL5902ACPZ-R2 ADL5902ACPZ-WP ADL5902-EVALZ
1
Temperature Range -40C to +125C -40C to +125C -40C to +125C
082008-A
SEATING PLANE
0.35 0.30 0.25
0.20 REF
COPLANARITY 0.08
FOR PROPER CONNECTION OF THE EXPOSED PAD, REFER TO THE PIN CONFIGURATION AND FUNCTION DESCRIPTIONS SECTION OF THIS DATA SHEET.
Package Description 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ] 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ] 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ] Evaluation Board
Package Option CP-16-10 CP-16-10 CP-16-10
Ordering Quantity 1,500 250 64
Z = RoHS Compliant Part.
Rev. 0 | Page 27 of 28
ADL5902 NOTES
(c)2010 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D08218-0-4/10(0)
Rev. 0 | Page 28 of 28

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